Performance Analysis of a 60 GHz Near Gigabit
System for WPAN Applications
L. Rakotondrainibe, Y. Kokar, G. Zaharia, G. Grunfelder, G. El Zein
European University of Brittany (UEB), INSA, IETR - UMR CNRS 6164 INSA, 20 Avenue des buttes de Coesmes, CS 70839 -35708, Rennes cedex, France firstname.lastname@example.org
Abstract—A 60 GHz wireless Gigabit Ethernet (G.E.) gain antennas. First of all, fading contributions are minimized communication system capable of near gigabit data rate has been by the spatial filtering effect of the antennas beamwidth, developed at IETR. The realized system covers 2 GHz available resulting in a higher coherence time. As shown in , when bandwidth. This paper describes the design and realization of the using directional antennas, the minimum observed coherence overall system including the baseband (BB), intermediate time was 32 ms (people walking at a speed of 1.7 m/s) which frequency (IF) and radiofrequency (RF) blocks. A differential is much higher than the lower limit of 1 ms (omnidirectionnal binary shift keying (DBPSK) modulation and a differential antennas). Then, the channel is considered invariant during the demodulation are adopted at IF. In the BB processing block, an coherence time and can be estimated once per few thousands original byte/frame synchronization technique is designed to of data symbols for Gbps transmission rate. Thus, the Doppler provide a small value of the preamble false alarm and missing effect (particularly due to the moving person) depends on the probabilities. For the system performances, two different real antenna beamwidth but it is not considered critical in indoor scenarios are investigated: measurements carried out in a large environments. The use of directional antennas also yield the gym and in hallways. Bit error rate (BER) measurements have benefits of reducing the number of multipath components (the been performed in different configurations: with/without RS (255, channel frequency selectivity) and therefore to simplify the 239) coding, with frame synchronization using 32/64 bits signal processing. As stated in , , the root mean square preambles. As shown by simulation, the 64 bits preamble (RMS) delay spread caused by multipath fading can be provides sufficient robustness and improves the system reduced to about 1 ns (the symbol duration for 1 Gbps with performance in term of BER. At a data rate of 875 Mbps, a BER -8BPSK modulation). This means that the channel coherence of 10 was measured at 30 m using high gain antennas for line-
of-sight (LOS) conditions. = 0.063/τ = 630 MHz bandwidth can be given as Bcoh, 0.9rmswhen using high gain antennas. In addition to a simple
differential demodulation (which offers higher tolerance to the Keywords-Millimeter-wave system; WPAN; single carrier; BER;
byte and frame synchronization inter-symbol interference (ISI) than others SC modulations),
the throughput less than 1 Gbps can be easily achieved I. INTRODUCTION without equalization. As the 60 GHz radio link operates only
in a single room configuration, an additional Radio over Fibre 60 GHZ wireless systems, currently under standardization (RoF) link is used to ensure the communications in all the within the unlicensed 57-66 GHz band, are aiming several rooms of a residential environment. For this reason, in this gigabits data rate for wireless personal area networks (WPANs) paper, we propose a hybrid optical/wireless system for the applications -. For any wireless system design, the indoor gigabit WPANs. The first system application in a selection of a modulation scheme is a main consideration and point-to-point configuration is the high-speed file transfer. has a large impact on the system complexity. In fact, problems Due to the cost of the transmission of the 60 GHz signals over such as power amplifier (PA) non-linearity and oscillator RoF, it is reasonable to transmit signals over the fiber at IF. phase noise are more important for these RF circuits resulting This paper is organized as follows. Section II describes the in performance degradation. These effects should be taken into transmitter (Tx) and the receiver (Rx). In this section, the account in the overall communication system. It was shown in baseband, the intermediate frequency and radiofrequency  that single carrier (SC) transmission has a lower tolerance blocks are presented. In Section III, measurement results are to phase noise and more resistant power PA non linearity than presented; this section represents the core contribution of the the multicarrier OFDM. Owing to these advantages, the paper. Section IV concludes the work. authors in  proposed the single carrier (SC) transmission for
multi-gigabit 60 GHz WPAN systems as defined in IEEE
II. TRANSMITTER AND RECEIVER DESIGN 802.15.3C standard. Up to now, in the literature, several
studies have considered propagation measurements , , Fig. 1 and Fig. 2 show the block diagram of the Tx and Rx potential applications, circuit design issues and several respectively. The multimedia data are transmitted from the modulations at 60 GHz -. However, few efforts have source (video server) through the G.E. interface of the 60 GHz been dedicated to the realization of a 60 GHz wireless system wireless transmitter. and its performance in a realistic environment.
Due to the high path loss at 60 GHz and the transmission
power restrictions, a simple solution is to use directional, high
This work is a part of Techim@ges research project supported by French “Media and Networks Cluster”, Comidom and Palmyre II projects financed by the “Region Bretagne”.
DML:Directly Modulated Laser (VCSEL)risk of packet loss since the source is always faster than the O/E:Optoelectronic converter (PIN photodiode )destination. In order to avoid the packet loss, a programmable DML logic circuit is used. The input byte stream is written into the O/E dual port FIFO memory (FPGA) at a high frequency 125 MHz. Source dataPattern ChannelPHY Preamble/The FIFO memory has been set up with two thresholds. When GeneratordataRS (255,239) encoder/the upper threshold is attained, the dual PHY block (controlled Scrambler/Differential encoder1.61 GHzRF 60 GHz by the FPGA) sends a „signal stop‟ (to the multimedia source) 8 807.43 MHz109.37 MHz reads out in order to stop the byte transfer. A slow frequency f100.92 MHz 1IF 3.5 GHz : 4X 3Multimedia sourceF2continuously the data stored in the FIFO. When the lower PLO 18.83 GHz PLO 3.5 GHz threshold is attained, the dual PHY block sends a „signal start‟ Gigabit Ethernet Clock managerinterface Txto launch a new Ethernet frame. Therefore, whatever the 70 MHz activity on the Ethernet access, the throughput at the output of 807.43 MHz the G.E. interface is constant. Then, the byte stream from the Files/Video serverG.E. interface is transferred in the BB-Tx, as shown in Fig. 4.
Figure 1. 60 GHz wireless Gigabit Ethernet transmitter FPGA Xilinx Virtex 4 TxSelect8RS (255,239)IF_TxPattern 8EncoderS/PIF 3.5 GHz 88generatorDual port88AGCDifferential 8ScramblerP/S FIFO memory8Encoder807.43 MbpsTsRF 60 GHz 875 MbpsEncoding X 3G.E. ControlinterfaceClockClock and data 70 MHz Clock managerfrecoveryf12109.37 MHz Channel 100.92 MHz 239 bytes239 bytes82391623916239816239168510807.43 MHz PLO 18.83 GHz Dataf= 100.928 MHz (source byte frequency) 1f= 109.375 MHz (channel byte frequency)2Synchro/ Descrambler/8 Gigabit Ethernet File transfer/RS (255, 239) decoderinterface Rxvideos streaming Figure 4. Transmitter baseband architecture (BB-Tx) Source data1.61 GHz BER A known pseudorandom sequence of 63 bits is completed Analyzer807.43 MHz with one more bit to obtain an 8 bytes preamble. This preamble
is sent at the beginning of each frame to achieve good frame
synchronization at the receiver. Due to the byte operation of RS Figure 2. 60 GHz wireless Gigabit Ethernet receiver coding, two clock frequencies f and f are used: 12The transmitted signal must contain timing information that FF (1) 12allows the clock recovery and the byte/frame synchronization f = = 100.929 MHz, f = = 109.375 MHz.1288at the receiver (Rx) . Thus, scrambling and preamble must 3.5 GHz2*239where: be considered. A differential encoder allows removing the F = = 875 MHz and F = F. 21242*(239+16)+8phase ambiguity at the Rx (by a differential demodulator).
As shown in Fig. 1, F is obtained from the IF. 2A. Transmitter design The frame format is realized as follows: the input source The Tx-G.E. interface is used to connect a home server to a byte stream is written into the dual port FIFO memory at a wireless link with about 800 Mbps bit rate, as shown in Fig. 3. slow frequency f. When the FIFO memory is half-full, the 1A header is inserted in the Ethernet frame to locate the starting encoding control reads out data stored in the register at a point of each received Ethernet frame at the receiver. higher frequency f. The encoding control generates an 8 bytes 2preamble at the beginning of each frame, which is bypassed by G.E-Tx interface 807.43 MHz/8the RS encoder and the scrambler. The RS encoder reads one GMIIFiles/ Video Rxbyte every clock period. After 239 clock periods, the encoding ServerData_validRJ45Txcontrol interrupts the bytes transfer during 16 clock periods, so J1FPGA Xilinx BB-TxDual Virtex 58DataIDE16 check bytes are added by the encoder. In all, two PHYXCV5LX20TRxRJ45successive data words of 239 bytes are coded before creating a ClockJ2TxFeedbacknew frame. After coding, the obtained data are scrambled link using an 8 bytes scrambling sequence. The scrambling
sequence is chosen in order to provide at the receiver the Figure 3. Transmitter Gigabit Ethernet interface lowest false detection of the preamble from the scrambled data.
Then, the obtained scrambled byte stream is differentially The gigabit media independent interface (GMII) is an interface encoded before the modulation. The differential encoder between the media access control (MAC) device and the PHY performs the delayed modulo-2 addition of the input data bit layer. The GMII is an 8-bit parallel interface synchronized at a (b) with the output bit (d): kkclock frequency of 125 MHz. However, this clock frequency is
different from the source byte frequency f = 807.43/8 =100.92 1 (2) d = db？k+1kkMHz generated by the clock manager in Fig. 1. Then, there is
correlator of 64 bits must analyze a 1-bit shifted sequence. The obtained data are used to modulate an IF carrier
Therefore, the preamble detection is performed with 64+7 = generated by a 3.5 GHz phase locked oscillator (PLO) with a 71 bits, due to the different possible shifts of a byte. In all, 70 MHz external reference. The IF signal is fed into a band-there are 8 correlators in each bank of correlators. pass filter (BPF) with 2 GHz bandwidth and transmitted through a 300 meters fibre link. This IF signal is used to FPGA Xilinx Virtex 4 Rxmodulate directly the current of a laser diode operating at 850 nm. At the receiver, the optical signal is converted to an 8888Dual portByte RS (255,239)8BER IF_RxP/SDescramblerS/PFIFO memoryalignmentDecoderanalyzerelectrical signal by a PIN diode and amplified. The overall
807.43 MbpsRoF link is designed to offer a gain of 0 dB. 875 Mbps
The IF signal is sent to the RF block. This block is 88Preamble G.E. composed of a mixer, a frequency tripler, a PLO at 18.83 GHz detectioninterfaceff21and a band-pass filter (59-61 GHz). The local oscillator frequency is obtained with an 18.83 GHz PLO with the same 82391623916851082391623916239 octets239 octets………….errors70 MHz reference and a frequency tripler. The phase noise of f= 100.928 MHz (source byte frequency) 1the 18.83 GHz PLO signal is about –110 dBc/Hz at 10 kHz off-f= 109.375 MHz (channel byte frequency) 2 carrier. The BPF prevents the spill-over into adjacent channels Figure 5. Receiver baseband architecture (BB-Rx) and removes out-of-band spurious signals caused by the 8 bitsmodulator operation. The 0 dBm obtained signal is fed into the ?00010000?64 bits 71 bitshorn antenna with a gain of 22.4 dBi and a half power Input 1Correlator C8 bitsdata 64 bits beamwidth (HPBW) of 10?V and 12?H. 2Correlator C64 bits 8Correlator CB. Receiver design ThresholdCorrelators-bank 0 & Register Periodicity ckeckThe receive antenna, identical to the transmit horn antenna, 64 bits 1Correlator Cis connected to a band-pass filter (59-61 GHz). The RF filtered 64 bits 2Correlator Csignal is down-converted to an IF signal centered at 3.5 GHz ?00010000?71 bits64 bits and fed into a band-pass filter with a bandwidth of 2 GHz. An 8Correlator C8 bitsautomatic gain control (AGC) with 20 dB dynamic range is used to ensure a quasi-constant signal level at the demodulator Correlators-bank 1input when the Tx-Rx distance or antenna moves. The AGC Unaligned loop consists of a variable gain amplifier, a power detector and data 15 bitsControl a circuitry using a baseband amplifier to deliver the AGC alignmentAligned Preamble voltage. This voltage is proportional to the power of the data Figure 6. The preamble detection and byte synchronization detection8 bitsreceived signal. A low noise amplifier (LNA) with a gain of
In addition, in order to improve the frame synchronization 40 dB is used to achieve sufficient gain. A simple differential performance, two banks of correlators are used, taking into demodulation enables the coded signal to be demodulated and
consideration the periodical repetition of the preamble: P (8 decoded. In fact, the demodulation, based on a mixer and a 1bytes) + D (510 bytes) + P (8 bytes) + D (510 bytes) + P (8 delay line (delay equal to the symbol duration Ts = 1.14 ns), 1223bytes). This process diminishes the false alarm probability (P) compares the signal phase of two consecutive symbols. A “1” fwhile the missing detection probability (P) is approximately mis represented as a ；-phase change and a “0” as no change, as multiplied by 2, as shown later. The preamble detection is in (2). Owing to the product of two consecutive symbols, the obtained if the same C correlators in each bank of correlators kratio between the main lobe and the side lobes of the channel indicate its presence. Therefore, the decision is made from 526 impulse response increases. This means that the differential successive bytes (P + D + P) of received data stored by the 112demodulation is more resistant to ISI effect compared to a receiving shift register. In fact, the value of each correlation is coherent demodulation. Nevertheless, this differential
compared to a threshold (;) to be determined. Setting the demodulation is less performing in additive white Gaussian
threshold at the maximum value (; = 64) is not practical, since noise (AWGN) channel. Following the loop, a low-pass filter
a bit error in the preamble due to the channel impairments leads (LPF) with 1.8 GHz cut-off frequency removes the high-
to a frame loss. A trade-off between P and P gives the frequency components of the obtained signal. For a reliable mfclock acquisition realized by the clock and data recovery threshold to be used. A false alarm is declared when the same
C correlators in each bank of correlators detect the presence of (CDR) circuit, long sequences of '0' or '1' must be avoided. kThus, the use of a scrambler (and descrambler) is necessary. the preamble within the scrambled data (D and D) . 12
The frame acquisition performance of the proposed 64 bits A block diagram of the baseband Rx is shown in Fig. 5.
preamble was evaluated by simulations and compared to that Owing to the RS (255, 239) decoder, the synchronized data
of the 32 bits preamble . The frame structure with 32 bits from the CDR output are converted into a byte stream. Fig. 6
preamble uses only a data word of 256 bytes (255 bytes + a shows the architecture of byte/frame synchronization using a
“dummy byte”). Fig. 7 and Fig. 8 show the missing probability 64 bits preamble. The preamble detection is based on the
versus channel error probability (p) and false alarm probability cross-correlation of 64 successive received bits and the internal 64 bits preamble. Further, each C(1 ? k ? 8) versus ; for an AWGN channel, respectively. k
Pm for 32 bits preamble0In these figures, P (or P) and P (or P) indicate the m1F1mF210missing (or false alarm) probabilities using one and two banks
of correlators, respectively. The effect of p on the false alarm -5probability is insignificant since the random data bits “0” and 10“1” are assumed to be equiprobable.
-3With the 64 bits preamble, for p = 10, the result indicate -10-10 -24 10that P = 10and P = 10for ; = 59. However, with the 32 mF2Pm1, Threshold = 29-7-13 Pm, Threshold = 29bits preamble, we obtain P = 10, P = 10 for ; = 29. This mF2Pm1, Threshold = 28 Pm, Threshold = 28means that, for a data rate about 1 Gbps, the preamble can be Pm1, Threshold = 27-15Miss detection probabilities10-7 Pm, Threshold = 27lost several times per second because P = 10 (; = 29) with m32 bits preamble. We can notice that, for given values of p and
P, the 64 bits preamble shows a smaller missing probability F2-20 10-4-3-2-110101010compared to that obtained with the 32 bits preamble. Channel error probability p After the synchronization, the descrambler performs the Pm for 64 bits preamble010modulo-2 addition between 8 successive received bytes and
the descrambling sequence of 8 bytes. At the receiver, the -510baseband processing block regenerates the transmitted byte
-10stream, which is then decoded by the RS decoder. The RS 10(255, 239) decoder can correct up to 8 erroneous bytes and -1510operates at a fast clock frequency f=109.37MHz. The byte 2 stream is written discontinuously into the dual port FIFO -2010memory at a fast clock frequency f. A slow clock frequency f21 Threshold = 58Threshold = 59= 100.92 MHz reads out continuously the byte stream stored -25Threshold = 6010by the register, since all redundant information is removed.
Afterwards, the byte stream is transferred to the receiver -3010Miss detection probabilities PmGigabit Ethernet interface, as shown in Fig. 9. The feedback -35 signal can be transmitted via a wired Ethernet connection or a 10-6-5-4-3-2-1101010101010Wi-Fi radio link due to its low throughput. Channel error probability p Figure 7. Miss detection probability with 32 bits and 64 bits preambles G.E-Rx interface807.43 MHz/8 Pf for 32 bits preamble, p = 1e-2GMII0101 corTxTo terminal2 corRJ45Data8RxJ3BB-RxFPGA Xilinx Dual IDEVirtex 5-5PHY10XCV5LX20TClockTxRJ45J4RxFeedback link -1010Figure 9. Receiver Gigabit Ethernet interface.
III. MEASUREMENT RESULTS -1510False alarm probabilities PfBack-to-back test of the realized system (without RF and AGC
loop) was firstly carried out. The goal is to evaluate the BER
-20of the realized system versus the signal to noise ratio (SNR) at 1005101520253035the demodulator input. An external AWGN is added to the IF Threshold modulated signal (before the IF-Rx band pass filter). The Pf for 64 bits preamble, p = 1e-2 010external AWGN is a thermal noise generated and amplified by 1 cor2 cor-5successive amplifiers. This noise feeds a band pass filter and a 10variable attenuator so that the SNR is varied by changing the -1010noise power. The BER versus SNR is shown in Fig. 10. -5Compared to the theoretical performance, for BER = 10, -1510the SNR degradation of the realized system is about 3.5 -2010and 3 dB for uncoded and coded data, respectively. This
degradation is mainly due to the 2 GHz available bandwidth. -2510This bandwidth is too wide for a throughput of 875 Mbps. In -30False alarm probabilities Pforder to avoid the increased power noise in the band, the filter 10
bandwidth must be reduced to 1.1 GHz (a roll-off factor 0.25) -3510. Using the free space model, Fig. 11 shows the estimated -40IF received power versus the Tx-Rx distance. This result takes 10010203040506070into account the transmitted power, the antenna gains, the path Threshold loss and the implementation losses of RF blocks. Two types of Figure 8. False alarm probability with 32 bits and 64 bits preambles
antennas were used: horn antenna and patch antenna. The Fig. 12, measured BER without BB blocks were obtained at
patch antenna has a gain of 8 dBi and a HPBW of 30?. The IF the CDR output for two possible configurations: AGC
receiver noise level is: minimum and maximum gains. In this case, the AGC loop is
disconnected but an external DC voltage controls the AGC -4 (3) N= -174 (dBm/Hz) + NF + 10log(B) = - 71.98 dBm, when the AGC Lamplifier. As a result, for BER around 10
amplifier is set at a gain of 8 dB, the upper limit of the Tx-Rx 9 where NF = 9 dB is the total noise figure and B = 2*10Hz is distance is about 7 m whereas this maximum distance can be the receiver bandwidth. As shown in Fig. 10, the minimum increased at 35 m, using an AGC amplifier with 28 dB gain. -4 SNR needed for BER = 10is about 10.5 dB. Thus, the The BER result with RS (255, 239) coding is also shown in receiver sensitivity is about S = - 61.5 dBm. Moreover, the Fig. 12 (frame structure using 32 bits preamble for a threshold demodulator input power must be greater than 0 dBm (due to -6; = 29). It is shown that for a BER at 10, the distance is equal the conversion loss of the power splitter and the minimum to 27 m without channel coding and 36 m with RS coding. power level required at the mixer input). These values give an This result proves the RS coding efficiency. Compared to the indication of the maximum distance accepted by the system. predicted distance obtained in Fig. 11, using high gain 010antennas, measured propagation loss characteristics showed DBPSK ideal without RS (255, 239)very good agreement. The effects of multipath propagation on DBPSK without RS (255, 239)-110DBPSK ideal with RS (255, 239)the BER performance are greatly reduced by the spatial DBPSK with RS (255, 239)-2filtering of the directive horn antennas. 10
-3-31010BER without RS (255, 239) & AGC loop disconnected (Gmin= 8 dB)BER without RS (255, 239) & AGC loop disconnected (Gmax= 28 dB)-4BER with RS (255, 239) & AGC loop disconnected (Gmax= 28 dB)10BER with RS (255, 239) & with AGC loop (Gmin to Gmax)-4BER10-510
-710BER -610-8 10468101214161820SNR (dB) -710
40Figure 10. BER versus SNR including AWGN channel
-8 10Power at IF-Rx, horn antenna Tx - horn antenna Rx510152025303540Distance (m)Power at demodulator input, horn antenna Tx - horn antenna Rx 20Power at IF-Rx, patch antenna Tx - horn antenna RxPower at demodulator input, horn antenna Tx - horn antenna RxFigure 12. BER versus the Tx-Rx distance (32 bits preamble) Power sensitivity at IF-Rx, for SNR = 10.5 dB (BER = 10e-4)Noise power at IF-Rx for NF = 9 dB0The main problem of using directional antennas is the human
obstruction. The signal reaching the receiver is randomly
affected by people moving in the area and can lead to frequent -20outages of the radio link. For properly aligned antennas, it is Minimum power level at the demodulator inputconfirmed that the communication is entirely interrupted when Power (dBm)-40the direct path is blocked by a human body (synchronization
loss during the obstruction duration). The measurement results Power sensitivity at IF-Rx
show a further attenuation of around 20 dB on the received -60power (as indicated in the power detector) when a person
moves in front of the receiver. High gain antennas are needed -80for the 60 GHz radio propagation but to overcome this major 0510152025303540Distance (m) problem, it is possible to exploit the angular diversity obtained
Figure 11. The IF received power versus Tx-Rx distance by switching antennas or by beamforming . To improve the
system reliability, a Tx mounted on the ceiling, preferably BER measurements were performed over distances placed in the middle of the room can mitigate the radio beam ranging from 1 to 40 meters in a large gym. At each distance, blockage caused by people or furniture , . In real the BER was recorded during 10 minutes. The Tx and Rx horn applications, the Tx antenna should have a large beamwidth to antennas (placed in the middle of the empty gym) were cover all the devices operating at 60 GHz in a room and the situated at a height of 1.35 m above the floor. These Rx antenna placed within the room should be directive so that measurements were conducted under LOS conditions with a the LOS components are amplified and the reflected fixed Rx and the Tx placed on a trolley pushed in a horizontal
components are attenuated by the antenna pattern. plane to various points about the environment. During the
In order to examine the effects of the antenna directivity measurements, the Tx and Rx were kept stationary, without
and the multipath fading, BER measurements were also movement of persons. A PN sequence of 127 bits provided by
conducted within hallway over distances ranging from 1 to 40 a pattern generator was used as data source. As shown in
meters, as shown in Fig. 13. The door of a 4 cm thickness events" and lead to permanent synchronization loss. Therefore, (agglomerated wood), was opened during the BER radio electric openings (windows ...) are necessary. measurements. The hallway has concrete walls and wooden CONCLUSIONS IVdoors on both sides. The Tx-Rx antennas (placed in the middle
of the hallway) were positioned at a height of 1.35 m. The idea This paper shows a full experimental implementation of a 60 was to analyze the results of BER measurements with and GHz unidirectional wireless system. The proposed system without RS coding in a wider hallway separated by a door. provides a good trade-off between performance and b) Top viewcomplexity. An original method used for the byte/frame a) Front viewTx2synchronization is also described. The numerical results show 4 m
that the proposed 64 bits preamble allows obtaining better Mobile Tx1BER results comparing to the previously proposed 32 bits 2.5 mtrolley2 mpreamble . This new frame format allows obtaining a high
preamble detection probability and a very small false alarm 2.84 mprobability. As a result, a Tx-Rx distance greater than 30
1.66 mmeters was reached with low BER using high gain antennas. Rx214.2 mOur investigation revealed that the high gain antenna Door separating Tx-Rxdirectivity stresses the importance of the antennas pointing HallwayStationary Dimensions: 60 m * 4 m * 2.5 m precision. In order to support a Gbps reliable transmission Rx1trolley(length * width * height)within a large room and severe multipath dispersion, a convenient solution is to use high gain antennas. However, the
use of directional antennas for 60 GHz WPAN applications is Figure 13. The hallway: a) front view; b) top view
very sensitive to objects blocking the LOS path. Due to the guided nature of the radio propagation along the Due to the hardware constraints, the first data rate was chosen hallway, the major part of the transmitted power is focused in at 875 Mbps. Using a new CDR circuit limited at 2.7 Gbps, a the direction of the receiver. However, the Rx horn antenna data rate of 1.75 Gbps can be achieved with the same DBPSK needs to be properly well aligned in the direction of the Tx, architecture or with DQPSK architecture. For suitable quality the misalignment of antennas (of a few degrees) results in a requirements in Gbps throughput, an adaptive equalizer should significant degradation of the LOS component and increases be added to counteract the ISI influence. the multipath components caused by the sides of walls and the
door borders. In a large gym, we found that the beam REFERENCES misalignment of directive antennas (of a few degrees) is less
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